Accurate control of peak and average current levels in switch mode power converters is critical to many power conversion applications. Peak current levels must not be exceeded to prevent saturation of the magnetic flux storage medium in the inductor, or magnetic flux storage element. Accurate control of the converter average output current is critical to protect load equipment during system failure modes, such as short circuit conditions, and to protect expensive power conversion elements from damage due to electrical stress. Input current control is needed to reduce the peak stress levels of electromagnetic interference (EMI) input filters components. Also, high power factor correction systems, such as switch mode power factor corrected rectifiers require accurate control of input current levels.
Power conversion systems often require wide bandwidth line and load response to maintain constant current levels into sensitive loads. Light Amplification by Stimulated Emission of Radiation (LASER) loads require precise current control because they respond exponentially to the stimulation current levels that are used to pump the LASER. Very short current stress levels result in catastrophic damage to the LASER, or degradation of the LASER's useable life. Loads, such as organic light emitting diodes, require large pre-charge current pulses, and benefit from being driven by high response rate power converters. Wide bandwidth power converters are needed in portable applications where large output filter capacitors cannot be used to meet load dump demands because they would require too much space.
Rapidly controlled transitions of the power converter output voltage are needed for many types of dynamic loads. This is the case for applications that require a variable output regulation point, such as the radio frequency (RF) power amplifiers in cell phones. The RF power amplifier often requires the highest operating power in cell phone applications. Supplying a variable level power source to the RF power amplifier can increase the RF power amplifier power efficiency significantly.
In RF power amplifier applications, it is important to be able to alter the power source output voltage rapidly, as well as maintain regulation with a minimal value output filter capacitor, if efficiency gains are to be realized. As the RF power amplifier supply is adjusted to meet the transmission power level needs for a specific cell phone environment, there is energy that is wasted in lost output filter capacitor charge. Thus, the power converter should have wide bandwidth response, as opposed to using large output filter capacitors to maintain regulation during rapid load current variations.
It is also critical to RF systems that the power converter have a controllable operating frequency. This is important to insure the switch mode power supply switching noise does not interfere with sensitive RF circuits.
Peak current mode control, average current mode control, and hysteretic average current mode control are typically used to meet some of the demands for power conversion as just described. All of the existing approaches have some adequate features but, unfortunately, these approaches also have many undesirable limitations.
Peak current mode control is typically used to address the needs of wide bandwidth voltage-to-current and voltage-to-voltage power conversion. Peak current mode control operates at a fixed frequency, and improves the bandwidth of the converter by regulating the peak inductor current, and thereby removing the effects of the inductor on the output filter response. Peak current mode control does not accurately regulate average current levels. Peak current mode control improves frequency response, but requires the addition of slope correction for duty cycles greater than 50 percent.
Average current mode control can also be used to meet some of the power conversion requirements as previously described. The operating frequency is fixed, and average current can be controlled accurately. Average current mode control does not require slope correction. However, average current mode control does not provide accurate control of peak current levels. Also, average current mode control uses a high gain current amplifier with ripple feed forward to improve line and load regulation, which can be susceptible to noise. Lastly, average current mode control requires continuously monitored current levels, and their associated losses.
Hysteretic average current mode control provides well regulated current levels and wide bandwidth response, does not require slope correction, but operates at a variable switching frequency, and requires continuously monitored inductor current levels with their associated power losses. A prior art example of a hysteretic average current mode power converter is shown in FIG. 1.
FIG. 1 is a schematic view of a prior art buck hysteretic average current mode converter 10. The buck mode voltage-to-voltage converter topology is presented in this example. However, the two state control method used in hysteretic average current mode control is applicable to many converter topologies, such as, but not limited to, boost, buck-boost, inverting, and active power factor corrected rectifiers.
The first terminal of an inductor 12 is selectively connected to input voltage source 34 with a charge switch 18, and selectively connected to GROUND with a transfer switch 20. The second terminal of the inductor 12 is connected to a filter capacitor 22 and load element 24. The charge switch 18 and transfer switch 20 are controlled by the state of the RS flip-flop (RSFF) 52, which provides the charge transfer control node 50. When the charge transfer control node 50 is in the logic high state, the charge switch 18 is on, and provides a low impedance connection of the inductor 12 between the input voltage source 34, and the filter capacitor 22 and load element 24. When the charge transfer control node 50 is in the logic low state, the charge switch 18 is off, and provides a high impedance disconnect. When the charge transfer control node 50 is in the logic low state, the transfer switch 20 is on, and provides a low impedance connection of the inductor 12 between the filter capacitor 22 and load element 24, and GROUND. When the charge transfer control node 50 is in the logic high state, the transfer switch 20 is off, and provides a high impedance disconnect. The charge switch 18 and transfer switch 20 are typically implemented as break before make switches, which prevents input voltage source 34 to GROUND shoot though currents during switch state transitions.
When the charge transfer control node 50 is in the logic high state, the charge switch 18 is on, and the transfer switch 20 is off. And, the input voltage source 34 is switch connected to the first terminal of the inductor 12. This provides a charging current path for the inductor 12 from the input voltage source 34 to the output node 36, which increases the inductor 12 current. When the charge transfer control node 50 is in the logic low state, the charge switch 18 is off and the transfer switch 20 is on. And, the first terminal of the inductor 12 is switch connected to GROUND through the transfer switch 20, and the second terminal of the inductor 12 is connected to the output node 36. This provides a discharging current path for the inductor 12. Thus, as the charge transfer control node 50 changes logic state, the power converter transitions between charging and discharging the inductor 12. This results in a ripple current as the inductor 12 current achieves a peak current level following the charging period, and a trough current level following the discharge period or transfer period.
The ripple current is monitored by the combination of a current sense element 32, current sense amplifier 14, and supporting passive components. This provides a voltage signal that is proportional to the inductor 12 current at the current monitor node 38. Continuous current monitoring power losses are often high in order to achieve reliable current sense signals, which decreases the power converter efficiency.
During the charging period the inductor 12 current increases, which causes the current monitor node 38 voltage to increase. The current monitor node 38 voltage is compared to the peak demand node 40 voltage by the peak comparator 46. When the current monitor node 38 voltage exceeds the peak demand node 40 voltage, the peak comparator 46 output transitions to the logic high state, and resets the RS flip-flop 52, causing the charge transfer control node 50 to transition to the logic low state. When the charge transfer control node 50 transitions to the logic low state, the charge period is terminated, and the transfer period begins.
During the transfer period the inductor 12 current decreases, which causes the current monitor node 38 voltage to decrease. The current monitor node 38 voltage is compared to the trough demand node 42 voltage by the trough comparator 48. When the current monitor node 38 voltage descends below the trough demand node 42 voltage, the trough comparator 48 output transitions the logic high state, and sets the RS flip-flop 52, causing the charge transfer control node 50 to transition to the logic high state. When the charge transfer control node 50 transitions to the logic high state, the transfer period is terminated, and the charge period begins.
The output node 36 voltage is regulated by controlling the average demand node 44 voltage at the output of the voltage sense amplifier 16 to null the difference between a scaled version of the output node 36 voltage and the voltage reference 35. When the output node 36 is below the desire output voltage level, the average demand node 44 voltage increases, which commands more current to the load element 24 and filter capacitor 22 to increase the voltage at the output node 36. When the output node 36 is above the desired output voltage level, the average demand node 44 voltage decreases, which commands less current to the load element 24 and filter capacitor 22 to decrease the voltage at the output node 36. Thus, the output is voltage regulated as the average demand node 44 voltage is adjusted in response to changes in the output node 36 voltage.
The average demand node 44 is connected to a summation element 28 and a difference element 30. The summation element 28 provides the peak demand node 40 signal by adding the average demand node 44 voltage to one half the ripple demand signal 26. The difference element 30 provides the trough demand node 42 signal by subtracting one half the ripple demand signal 26 from the average demand node 44 voltage. The ripple demand signal 26 is scaled by one half by the multiplication element 54. The ripple demand signal 26 is set to control the ripple current level of the converter, which is proportionally related to the converter switching frequency. However, the converter switching frequency is not accurately controlled, and varies with input voltage source 34, output node 36 voltage, and inductor 12 value. Converter switch frequency variation is highly undesirable for sensitive RF and analog signal condition applications. Therefore, hysteretic average current mode control does not meet the requirements for frequency regulation that are needed for many power conversion applications.
FIG. 2 is a waveform view of prior art signals for the hysteretic average current mode power converter depicted in FIG. 1. This figure illustrates how inductor 12 current, which is proportional to the current monitor node signal 62 voltage, increases and decreases as the peak demand node level 56 and trough demand node level 58 are exceeded and descended below respectively. The average demand node level 60 is centered at one half the ripple demand signal level 70 from the peak demand node level 56 and trough demand node level 58. The charge slope 72 can be calculated by the charge slope equation 64. The transfer slope can be calculated by the transfer slope equation 66. The charge slope equation 64 and transfer slope equation 66 are accurate approximations that are well accepted, and used by power converter designers for deigning switch mode power converters. The switching frequency can be calculated from the charge slope equation 64 and the transfer slope equation 66, and the ripple demand signal level. The switching frequency equation 68 illustrates that conventional hysteretic average current mode control is inversely proportional to inductor 12 ripple current and inductor 12 inductance, and also varies with the input voltage source 34 (Vin) and the output voltage node (Vout).
As illustrated by the description of the prior art, hysteretic average current mode control controls peak, average, and minimum current levels, and achieves wide bandwidth operation without the need for slope correction. However, operating frequency varies with the power converter inductor value, Vin and Vout, and power losses are increased due to continuously monitored inductor current levels.